With both current and voltage feedback operational amplifiers (op amps) available to the system designer, how do you select which device to use? This two-part article discusses applications most suited to each type of op amp, and why certain applications are unsuitable for one or the other type of amplifier. Widely-used circuits are shown with example designs. The emerging fully differential amplifier type of amplifier (a special case of a voltage feedback device) is also shown along with several suitable applications. **Part 1** reviews the internal differences between the two types of op amps and presents some key applications most suitable to voltage feedback type devices.

**Initial amplifier selection criteria**

There is a bewildering array of possible op amps for a designer to select from in any given application. However, there are distinct internal differences between the two major types of amplifiers: voltage feedback (VFB) and current feedback (CFB). These differences may lead a designer to choose one over the other in certain applications. A newer type of op amp, the fully differential amplifier (FDA), is a type of voltage-feedback op amp that includes an output common-mode control loop. It has the same application emphasis as a standard voltage feedback amplifier, as well as several specific applications ideally suited to this type of amplifier.

Before giving any attention to the type of amplifier to use, most designers first should consider the output signal requirements (for example, maximum desired output V_{pp} and output current, as well as load type) and necessary dynamic characteristics such as settling time, full power bandwidth, or a distortion requirement. Using these specifications, the universe of possible amplifiers can be narrowed to devices that can handle the correct range of supply voltages needed to deliver the output V_{pp} , and then limited further to a minimum supply current versus desired output dynamic characteristic. Normally, a CFB gives the best power efficiency for higher-frequency dynamic range needs, while a VFB generally has better noise and/or DC precision, and is the part of choice for certain types of circuits.

**Internal comparison of voltage and current feedback op amps**

To understand why some circuits work better with one or the other type of amplifier, you need to first understand the internal topology of each amplifier and their resulting transfer functions. Compare a simplified expression for the Laplace transfer function written in loop gain format. **Figure 1** shows the internal block diagram of a VFB along with the internal model and closed loop transfer function. A(s) is the frequency-dependent open-loop gain of the op amp. It is modeled here as a single dominant pole response. Figure 1 uses an inverting gain configuration for comparison purposes because the FDA device is most clearly understood as a differential inverting configuration.

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*Figure 1. Open loop to closed loop conversion for voltage feedback.*

The transfer function has the desired gain in the numerator (-R_{F} /R_{G} ) and a loop gain (LG) term in the denominator that determines the frequency response. One way to understand this LG is to plot [20 · Log (A(s))] and [20 · Log(1+R_{F} /R_{G} )] on the same grid. The key issues for this plot are: 1) the separation between these two curves at lower frequencies (this separation shows the magnitude of the loop gain); and 2) at what frequency they intersect.

At the point of intersection, sometimes called loop gain crossover, the term in the denominator of the transfer function drops to 1 + 1e^{-j?} (where j? is the angle of that expression). The important check is that this angle is well away from -180 degrees to avoid closed loop oscillations or peaking. **Figure 2** shows an example loop-gain plot of these two terms, where a simple single-pole response for A(s) is assumed and no phase added by the feedback network.

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*Figure 2. Loop gain and phase for a VFB op amp. X-axis is log frequency.*

In Figure 2, the (1 + R_{F} /R_{G} ) term is assumed to add no phase impact to the loop gain; only the open-loop phase of A(s) introduces loop phase shift in this simple example. This graph is the lower plot, where the loop gain crossover frequency is mapped down to find the remaining phase margin at that frequency.

Note that it is impossible for the VFB to change the signal gain without changing the loop-gain characteristic. This effect is where the gain bandwidth product (GBP) concept arises. If the gain increases, the bandwidth must decrease. If the gain decreases, the bandwidth increases, and the phase margin will normally decrease. (See **Reference 1** for a more complete discussion.)

In contrast, the internal workings of a CFB op amp are quite different. Those blocks and the resulting closed loop transfer function for the inverting configuration are shown in **Figure 3** where, again, an inverting configuration is used and a single pole internal transimpedance gain is assumed for Z(s).

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*Figure 3. Current feedback internal structure and closed loop response.*

The CFB uses a unity-gain buffer across the two inputs. This forces the inverting node voltage to follow the non-inverting input voltage. That buffer is intended to present a low impedance to the inverting port, where a low level error current may be sensed and passed on to the output through a transimpedance gain. It is this internal transimpedance gain, Z(s), which acts in the same fashion as the VFB A(s) to provide a high DC gain with a dominant pole.

When the loop is closed, the same desired gain is achieved; but the loop-gain terms are very different. The CFB amplifier has a loop gain set by the forward transimpedance gain compared to the feedback impedance. **Figure 4** plots the loop gain and phase for a typical CFB amplifier, where the feedback element is assumed to introduce no phase shift in this simplified analysis.

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*Figure 4. Current feedback (CFB) loop gain and phase plots. X-axis is log frequency.*

This plot looks very similar to the VFB plot except that the external element setting the loop gain is the feedback impedance alone. The greatest difference between the VFB and CFB amplifier is that the loop gain can be set separately from the signal gain using the feedback impedance. The feedback impedance becomes an independent compensation element, where the gain can then be set using the normal gain equations from whatever impedance value is selected for R_{F} . This approach gives what is sometimes called *gain bandwidth independence* for the CFB amplifier. (See Reference 1 for a more detailed discussion.)

The final type of amplifier to be considered here is the new fully differential amplifier (FDA). **Figure 5** shows the configuration and closed loop transfer function for this type of amplifier.

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*Figure 5. Fully differential amplifier (FDA) structure and transfer function.*

If the two feedback networks are allowed to be unmatched, the transfer function is fairly complicated. If they have a matched divider ratio (see Figure 5), the equations simplify to be the same as the inverting VFB transfer function. The effect of the separate common-mode loop is not shown. This loop acts to servo the average output voltage to a value set by a V_{OCM} input pin voltage (see **Reference 2** for a more complete discussion of the FDA topology). For applications considered in this article, the FDA is treated as differential VFB device.

**Application imperatives**

In the range of possible applications for wideband op amps, several types must use VFB devices. These circuits can sometimes be forced to work using CFB devices, but usually at the cost of complexity and poorer performance. Any circuit that requires flexibility in the feedback element and/or capacitors in the feedback will have stability problems with a CFB device. This instability is the result of the loop gain depending on the feedback impedance. Therefore, any circuit that needs a lot of flexibility in that impedance is going to interact with the achievable frequency response if a CFB amplifier is used.

The following example circuits should use a VFB device for implementation.

**A. Transimpedance amplifiers** . These circuits take a current source input, typically from a capacitive source, and turn it into a voltage at the output. The feedback resistor is the gain element and normally needs a compensation capacitor in parallel for correct operation. **Figure 6** shows an example using the OPA657, a very wideband JFET input device uniquely suited to the transimpedance application. This device is a non-unity gain stable VFB with relatively low input noise voltage and very high gain bandwidth product. For a given diode source capacitance, the amplifier gain bandwidth product (GBP) determines the achievable bandwidth and/or transimpedance gain (**Reference 3** ).

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*Figure 6. Example transimpedance design using the OPA657.*

In this example, the 500 kO along with the 200 pF diode capacitance gives a noise gain zero at approximately 1.6 kHz. With the feedback capacitor set to achieve a maximally flat Butterworth response, the resulting F_{-3dB} will be at the geometric mean of this zero and the 1.6 GHz gain bandwidth product of the OPA657. (Reference 3 gives a detailed analysis for compensation and noise in a transimpedance design.) **Figure 7** shows the 1.6 MHz transimpedance bandwidth in a simulated frequency response of Figure 6.

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*Figure 7. Simulated transimpedance frequency response of Figure 6.*

**B. Integrator based circuits** . These applications are looking for a capacitive feedback to implement an integrator function. A good example would be the Multiple feedback (MFB) active filter. **Figure 8** shows an example using the OPA820, a low-noise, wideband voltage feedback op amp uniquely suited to this application.

Embedded within this filter circuit is an integrator configuration that arises from the 200 O and the 12.5 pF feedback capacitor.

At very high frequencies, that capacitor shorts out, giving a local unity-gain feedback for the amplifier. This result suggests that a unity gain stable VFB should be used in this type of circuit to avoid high frequency oscillations. This particular example is targeting a 10 MHz low-pass Butterworth response with an in-band gain of -4 V/V. (**Reference 4** describes how to choose the R's and C's in the MFB filter for improved noise and distortion.)

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*Figure 8. Example MFB filter using the OPA820.*

Since the local feedback capacitor shorts out at higher frequencies, this circuit normally needs to use a unity gain stable VFB. (**Reference 4** suggests a path to improved phase margin in this circuit using non-unity gain stable amplifiers.) One of the advantages of the MFB design is improved stopband rejection (over the Sallen-Key filter), and very good filter accuracy versus finite amplifier gain bandwidth product. This filter tends to achieve a slightly lower Q than that targeted as the O_{0} moves closer to the amplifier GBP.

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*Figure 9. Simulated 10 MHz Butterworth filter of Figure 8.*

**C. Circuits with simple feedback poles implemented as an RC in the feedback network** . Designers often implement a low-pass pole in this fashion. Since the feedback impedance is now a parallel RC network, this circuit must be implemented with a VFB, and preferably a unity-gain stable VFB. This type of simple filtering works best for inverting signal paths. In that case, the input signal sees the gain resistor as an input impedance that converts the signal to a current into the inverting node. From there, it continues into the feedback impedance to set the gain to the output.

This type of pole implemented for a noninverting configuration has a pole/zero pair because the gain transitions from a DC value set by the resistors to unity gain as the feedback capacitor shorts out at higher frequencies. Therefore, this circuit drops to unity gain if implemented as a non-inverting stage. In the inverting configuration, the gain continues down with a one pole response. It is important to notice that a unity-gain stable amplifier would be preferred because the noise gain drops to unity for either the inverting or non-inverting application.

**Figure 10** shows an example using the THS4281 (a low-power, rail-to-rail output VFB) in the inverting configuration. In this example, the signal path was AC-coupled through a blocking capacitor. This allowed the non-inverting bias voltage to be set to mid-supply and then have a DC gain of one to the output. Because the THS4281 includes the negative rail on the input, an alternative DC-coupled design is possible by changing the lower 8 kO resistor on the non-inverting input to 889 O and removing the input blocking capacitor.

This AC-coupled design allows a direct comparison of this circuit to a low-power CFB that does not have an input range extending to ground. That device, the OPA684, also uses less than 2 mA supply current and gives greater than 100 MHz bandwidth in most applications. **Figure 11** compares the simulated response of this 1 MHz one pole rolloff circuit of Figure 10, using both the THS4281 and the OPA684 (where the OPA684 would use a single +5 V supply). Both parts give the expected one pole rolloff, but the OPA684 shows a very anomalous response at higher frequencies indicative of probable oscillations.

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*Figure 10. Inverting band-limited design using the THS4281.*

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*Figure 11. Simulated inverting low pass filter design.*

**D. Wideband DC-coupled amplifiers with good DC precision** . VFB op amps have better input offset voltage and (for bipolar amplifiers) matched input-bias currents. As a result, considerably lower output DC-offset voltage and drift can be delivered as compared to equivalent CFB implementations. This consideration becomes more important as the required gain increases. **Figure 12** shows an example high gain, DC-coupled inverting amplifier stage using a non-unity gain stable VFB. The OPA846 offers very good input offset voltage and offset current along with a low 1.2 nV/vHz input voltage noise.

To take advantage of this low noise, the gain of -20 circuit in Figure 12 implements a matched 50 O input impedance using only the input resistor and then a 1 kO feedback resistor. If a 50 O DC-coupled source impedance is assumed, the required bias current cancellation resistor on the non-inverting input would be 100 O| in parallel with 1 kO = 90.9 O. Decouple this resistor with a parallel capacitor to reduce its high-frequency noise contribution.

Another interesting aspect of this circuit is the noise gain (NG), which is also the gain for the input offset voltage, is reduced below the inverting signal gain because of the 50 O source resistor. Including that value in the NG equation gives an NG = 11 while the signal gain will be -20.

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*Figure 12. Inverting gain of -20, low-output V*

_{os}design, using the OPA846. This high-gain inverting implementation using the OPA846 can also be done using a low-power current feedback amplifier. For comparison, an OPA684 low-power CFB is compared in **Figure 13** for the simulated frequency response. Since high equivalent gain bandwidth is a natural advantage for CFB amplifiers, the OPA684 gives very similar small signal bandwidth to the OPA846 of approximately 150 MHz. The OPA684 uses 1.8 mA quiescent current while the OPA846 uses 12.9 mA.

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*Figure 13. Gain of -20V/V simulated performance using VFB and CFB devices.*

A calculation of the maximum output DC offset error band at 25 °C for the OPA684 shows the relatively poor DC precision offered by the CFB implementation. That calculation must treat the two input bias currents separately, because they are not physically related in the CFB input stage as they are in most VFB input stages. Leaving the 90.9 O resistor on the non-inverting input, the total output DC error band, using the OPA684 specified maximum DC errors at 25 °C, is:

Maximum V_{os} =

±3.5 mV·11 V/V±10µA·91 O·11 V/V± 16µA·1 kO =

±64.5 mV

So, while the OPA684 can match the speed of the OPA846 in this higher gain application, if output DC precision (or lower noise) is desired, the VFB design offers considerable advantages.

**Conclusions to Part 1**

In Part 1, we reviewed the key internal differences between VFB and CFB op amps as a means of identifying applications that must use the VFB topology over the CFB devices. These examples generally fall into two categories: applications that need a capacitive feedback for some reason; and circuits that need to emphasize DC precision and/or the lowest output noise. **Part 2** will continue with applications uniquely suited to the CFB topology, and then introduce the newest member of the wideband op amp universe—the fully differential amplifier. Part 2 will conclude with illustrations of a few applications uniquely suited to this amplifier.

*(To go to Part 2 of this article, click here)*

**About the author**

*Michael Steffes* * is the Market Development Manager for High-Speed Signal Conditioning, and a Distinguished Member of the Technical Staff, at Texas Instruments Inc. With more than 25 years of experience in high-speed amplifier design, applications, and marketing, Michael currently provides product definition and customer design-in support. *

Michael earned a BSEE from the University of Kansas and an MBA from Colorado State University. He shares several basic patents in high-speed op amp designs and has written more than 85 product data sheets, scores of contributed articles, applications notes and conference papers. You can reach Michael at ti8728499steffes@list.ti.com.

**References**

1. “Voltage Feedback vs. Current Feedback Op Amps,” Jim Karki, Texas Instruments Application Note, SLVA051 (Voltage Feedback vs. Current Feedback Op Amps or click here).

2. “Fully Differential Amplifiers,” Jim Karki , Texas Instruments Application Note SLOA054D (Fully-Differential Amplifiers (Rev. D) or click here).

3. “Control Frequency Response and Noise in Broadband, Photodetector Transimpedance Amplifiers,” Michael Steffes, EDN, Design Feature, July 4th, 1996 pp. 113-125.

4. “Design Methodology for MFB Filters in ADC Interface Applications,” Michael Steffes. TI Application Note, SBOA114 (Design Methodology for MFB Filters in ADC Interface Applications or click here).

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