In my previous blog in this series, we looked at the video amplifier used in a piece of professional video equipment, the Sony BE-3000 video switcher/effects board. Figure 1 shows a video amplifier similar to one in the Ampex AVR-3 video tape recorder's demodulator board. In the original circuit, Q4's collector was connected to the feedback resistor directly, and the variable capacitor CF was not needed. A separate buffer amplifier then provided drive via the collector of Q4.
This amplifier has a slightly different second voltage gain stage that allows more symmetrical output voltage swings by tying the collector of Q4's load resistor to the minus rail. In this example, the amplifier is set for a gain of four instead of two. And for optimal flatness in video frequency response, a variable capacitor (CF) is adjusted. Amplifiers similar to this have been seen in many audio devices.
The measured open loop gain and -3 dB open loop frequency were about 514 V/V and 560 kHz, respectively. The gain bandwidth product is about 288 MHz.
For a quick calculation of the open loop frequency response, first we determine the input resistance into Q4's base. The internal base emitter resistance is r•πQ4 ~175Ω given the DC collector current of Q4 is about 15 mA and with β = 100. Therefore, the input resistance at the base of Q4 is about
This input resistance is in parallel to RL_Q1 (402Ω), and the paralleled resistance is 1185‖402Ω = 300Ω.
To determine the open loop pole, we calculate the input capacitance to the base of Q4 via the voltage gain of Q4 and the Miller multiplier capacitor Cc_Miller.
The voltage gain (A) of Q4 is about 820/[(1/gmQ4 ) + 10] ≈ 820/[1.73 + 10] = 70 V/V. There is an internal collector-base capacitance CcbQ4 of about 2.5 pf.
Thus the equivalent capacitance at the base of Q4 is approximately:
(1 + A)(Cc_Miller + CcbQ4 ) = (1 + 70 )(10 + 2.5) = 887.5pf. The calculated open loop pole then 1/[2π(300Ω)(887.5pf)] = 598 kHz.
With a closed loop gain of +4 V/V, this amplifier's frequency response was well within 0.2 dB out to 4.2 MHz. The differential phase and gain were about 0.15° and 0.1 percent, respectively.
One may ask about comparing it with op-amps of the 1970s or 1980s. For the comparison, an LM318 was chosen, since its high slew rate (50 V/μs) and gain bandwidth product (50 MHz) would be suitable for video use.
The LM318 was set for a closed loop gain of 2 V/V by using 1000Ω resistors for the feedback network. Unfortunately, the differential phase and gain measured about 10° and 5 percent, respectively.
See Figures 2-5 for a comparison between the input and output signals of the LM318 and the two op-amps we described in detail in our previous blog. As seen in Figure 3, the LM318 has an uneven frequency response (e.g., a peak of nearly 1 dB) that also causes overshoot.
Note a slightly peaked frequency response for this 50 MHz gain bandwidth product op-amp.
The amplifier circuit from our previous blog's Figure 2 also had a 50 MHz gain bandwidth product and exhibited much flatter frequency response than the LM318.
With a gain of four (instead of two), this amplifier from Figure 1 still provides a very flat frequency response.
By the 1990s, current mode feedback amplifiers such as the EL2120 and EL5261 provided superior video frequency response and very low differential phase and differential gain distortion. Likewise, voltage feedback amplifiers such as the ISL55002 will perform superbly in terms of video specifications.