SLIC Supplies

Subscriber-line interface cards (SLICs)
provide the interface between the telephone service provider and
the telephone handset in your home. They operate in two main modes:
on-hook refers to when the handset is idle and waiting for a signal
that indicates someone wants to make a connection, and off-hook
refers to when the handset is active and the user is trying to
complete a connection. Telephone-system voltages are traditionally
negative to prevent electromigration from eroding the installed
copper wiring.

Telephone systems require certain special voltages that vary
from application to application and country to country. This
article presents circuits for deriving these voltages from commonly
available supply voltages. Table 1 summarizes the input and output
characteristics of the circuits.

Circuit Input Output 1 Output 2 Output 3
Figure 1 +4.5 to +5.5 V -48 V @ 300 mA
Figure 2 +5.5 V minimum input provided by a wall adapter -24 V @ 150 mA -100 V @ 50 mA
Figure 3 +12 V -24 V @ 50 mA -48 V @ 100 mA
Figure 4 +12 V -24 V @ 400 mA ±5% -72 V @ 100 mA ±5%
Figure 5 +5 V Isolated +3.3 V @ 100 mA Isolated -24 V @ 100 mA Isolated -72 V @ 25 mA

Table 1 : Inputs and outputs.

The on-hook voltage, which generates the ringer voltage, is
typically -72 V in the United States and as high as -150 V in other
countries. The ringer voltage, a 20- to 60-Hz sinusoid, drives an
electromechanical bell in the handset that can be located far from
the central office (CO). The off-hook voltage is typically -48 V in
the United States, although some localized systems use -24 V. This
voltage powers the system during voice communications. The CO
provides power for the telephone system independently from the
electric utility, allowing the telephone to work during a power

With advances in data-communication technology, companies are
incorporating voice service with data service to provide integrated
communication systems. Such systems require SLIC functions to
maintain compatibility with legacy equipment. The following
circuits demonstrate techniques for generating SLIC voltages from
commonly available voltages. All of the circuits are based on a
transformer flyback topology using a MAX668 boost controller. This
topology achieves compact magnetics and flexible output

-48-V Output from +5-V Supply

The circuit in Figure 1 generates -48
V at 300 mA for customer premises equipment (CPE) or client-side
equipment from a +4.5-V minimum input. The input voltage also is
the gate-drive voltage for the MAX668 (U1), limiting the input
voltage to +5.5 V (max). The MOSFET switch (Q1) presents more gate
capacitance than the controller can drive efficiently; therefore,
complementary emitter followers (Q2, Q3) buffer the gate-drive

Q1 is selected to switch 9-A peak current with VGS = 3.8 V (4.5
VIN-VBE). A snubber for the flyback voltage is not necessary
because the breakdown voltage of Q1 is nine times the voltage
reflected back from the secondary to primary transformer winding.
R7 and C7 filter the current-sense signal to prevent
false-triggering caused by switching noise. The moderate switching
frequency (165 kHz) allows good efficiency with moderate-cost and
moderate-performance parts.

The transformer (T1) is wound on a Coiltronics SG4 gapped core
with AL = 75 nH/T². The primary winding is eight turns of
#22AWG, so the primary inductance is 4.8 µH. The secondary
winding is 64 turns of #28AWG, so the turns ratio is 1:8. The
switch duty cycle is approximately 55%, which gives close to the
optimal power transfer for a given magnetic volume.

A lower cost, fast-recovery diode (D1) is used instead of a
Schottky diode because the lower switching frequency minimizes the
impact of the reverse-recovery time on the switching efficiency.
Also, the high-output voltage minimizes the advantage of a lower
forward-voltage diode. Because the output voltage is negative, the
feedback must be inverted by an op amp (U2) to match the switching
controller (U1). D2 protects the inverting input from being pulled
negative. R3 and C5 provide the dominant pole for feedback-loop

-24- and -100-V Output from +5.5-V

The circuit in Figure 2 generates -24
V at 150 mA and -100 V at 50 mA for CPE from a +5.5-V minimum input
provided by a wall adapter. The MAX668 (U1) has an internal linear
regulator that generates a +5-V rail for the gate-drive voltage.
The MOSFET (Q1) is selected to switch 7-A peak current with VGS =
4.5 V. Snubbing the flyback voltage is not necessary because the
leakage inductance is low and the breakdown voltage of Q1 is more
than two times the reflected output voltage. R5 in series with the
gate of Q1 slows down the turn-on time to minimize the switching
noise seen by the current-sense amp.

The transformer (T1) is an off-the-shelf unit from Coiltronics
that allows fast circuit development. A custom transformer can be
designed and optimized for volume production. The primary
inductance is 3.8 µH, and the turns ratio is 1:1:3. This makes
the duty cycle approximately 80% instead of 50% for optimal power
transfer. The result is higher peak current compared with an
optimized transformer.

The -24-V output is fed back through the op amp inverter (U2),
regulating the output to ±1% directly. The -100-V output uses
the transformer-turns ratio for regulation. This works as long as
the power output from the -100-V supply is not significantly more
than the power output from the fed-back voltage. The typical
application does not require tight regulation of the -100-V on-hook
voltage, and so ±10% is sufficient.

-24- and -48-V Output from +12-V

The circuit of Figure 3 generates -24
V at 50 mA and -48 V at 100 mA from a +12-V nominal input. It
demonstrates the optimal turns ratio for power transfer in the
transformer. The primary to -24-V secondary turn ratio is 1:2, and
the primary to -48-V secondary turns ratio is 1:4. Therefore, the
switching regulator operates at 50% duty cycle. The -48-V output is
fed back to the controller for regulation. The -24-V output is
regulated to ±5% by the turns ratio and close-coupling of the
secondary windings.

-24- and -72-V Output with Split
Feedback from +12-V Supply

The circuit in Figure 4 generates -24
V at 400 mA and -72 V at 100 mA from a +12-V nominal input. Both
outputs are regulated to ±5% under all combinations of line
and load by splitting the feedback between the two outputs. The
trade-off is to give up a little tolerance on the off-hook voltage
for tighter tolerance on the on-hook voltage. The feedback ratios
are based on the relative output powers. The -24-V output delivers
4/7th of the maximum output power, so the feedback resistor is
scaled to supply 4/7th of the current required for regulation.
Similarly, the -72-V output delivers 3/7th of the maximum output
power, so the feedback resistor supplies 3/7th of the current
required for regulaton.

The transformer is a custom design for this application. The
leakage inductance, which represents the imperfect coupling between
the primary and secondary windings, is large enough to require
snubbing of the primary flyback voltage to prevent breakdown of the
switching transistor. R8 and C8 slow down the switch transition and
dissipate some of the energy in the leakage inductance to limit the
maximum flyback voltage. The Coiltronics Versa-Pac line of
transformers are tri-filar wound for maximum coupling (i.e., three
wires are wound in parallel), so leakage inductance is minimized.
The trade-off is decreased flexibility in the turns ratio and lower
isolation-voltage rating between the primary and secondary

Isolated +3.3-, -24-, and -72-V Output
from +5-V Supply

The circuit of Figure 5 generates
isolated +3.3 V at 100 mA, -24 V at 100 mA, and -72 V at 25 mA from
a nominal +5-V input. Isolation is required when the input voltage
is not isolated from the line (either electric utility or telephone
system power). The transformer isolates the +5-V input from the
output voltages. The +3.3-V output is generated from an extra
secondary winding. A linear post-regulator is required because of
the wide voltage ratio between +3.3 and -24 V used for feedback.
The -72-V output regulation is not critical and relies on the turns
ratio and close-coupling to the -24-V secondary winding.

The isolated feedback from the -24-V output is implemented with
a shunt regulator and an optoisolator. The shunt regulator combines
a voltage reference and an error amplifier to generate a
current-error signal. The error current drives the photodiode in
the optoisolator, which modulates and isolates the current in the

The optoisolator is selected for nominally 100%
current-transfer-ratio at 10 mA. The error current is converted to
an error voltage through R4. R4 and C13 create a pole in the loop
response that limits the loop bandwidth to 2.8 kHz. The loop
compensation must take into account the signal delay of the
optoisolator as well as the additional gain of the shunt regulator
combined with the error amp in the MAX668.

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