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Using high-side current-sense amplifiers with input series resistors

Current-sense amplifiers are extremely useful for high-side current-sensing applications that need to amplify small voltages across a sense resistor on a high-voltage rail, and feed it to a low-voltage A/D converter or a low-voltage analog control loop. For example, you may need to measure the current flowing in a circuit or to a load using a low-ohm sense resistor. In these applications the current-sense signal frequently needs to be filtered at the source, such as across the sense resistor.

When discussing functional operation, a current-sense amplifier can be considered an instrumentation/differential amplifier with a floating input stage. This means that even when the device is powered from a single-supply with VCC = 3.3 or 5 volts, it can amplify input differential signals at a common-mode voltage well beyond these power supply rails. The common-mode voltages in a current-sense amplifier can, for example, be up to 28 V (MAX4372 and MAX4173) and 76 V (MAX4080 and MAX4081).

The design could use either a differential filter (Figure 1 ) to smooth “spiky” load currents and sense voltages:


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Figure 1: Diagram for a differential filter to smooth spiky load currents.

or a common-mode filter (Figure 2 ) to enhance ESD operation/immunity to common-mode voltage spikes and temporary overvoltages:


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Figure 2: Circuit diagram for a common-mode filter to improve immunity to ESD spikes and common-mode overvoltages.

These filters can be successfully implemented by choosing the right component values. If the wrong component values are selected, however, unplanned input offset voltages and gain errors can be introduced, which compromise circuit performance.

Determining Which Filters to Use
Consider the MAX4173 current-sense amplifier shown in Figure 3 .


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Figure 3: Internal functional diagram of the MAX4173.

This device has its sense resistor connected directly to the RS+ and RS- terminals of the chip. The differential voltage across the sense resistor is reproduced across RG1 by internal operational-amplifier function so that ILOAD x RSENSE = VSENSE = IRG1 × RG1 . This current (IRG1 ) is then level-shifted and amplified by an internal current mirror to generate the output current, IRGD . The internal circuit for the MAX4173 implements RGD = 12 kO and RG1 = 6 kO.

Therefore,

VOUT = RGD × IRGD

= RGD × Gain × IRG1
= RGD × Gain × VSENSE / RG1

As RGD and RG1 are on-chip resistors, their actual values normally vary by as much as 30% due to semiconductor process variations. However, because it is the ratio of RGD and RG1 that determines the final gain accuracy, the final gain is well controlled and can be easily trimmed during manufacture.

However, when series resistors are inserted between the RSENSE+ and RSENSE- terminals of a sense resistor, and RS+ and RS- pins of the part to implement differential/common-mode filters (as shown in Figure 1 and Figure 2), the chip behaves as though RG1 and RG2 have been modified. From the above equation, it is apparent that modifying a trimmed RG1 introduces a gain error. Further, since the absolute value of RG1 can vary by as much as ±30%, this gain error can also vary by ±30% and cannot be controlled or predicted between multiple parts. The only way to control this gain error is, therefore, to ensure that the input series resistor, RSERIES+ , is small compared to RG1 .

Additionally, a mismatch between resistors RG1 and RG2 is “converted” by the device's input bias currents into an input offset voltage. The MAX4173 and MAX4372 data sheets show that bias current IRS – is twice IRS+ , and therefore, any resistor in series with RG1 (RSERIES+ ) should be twice that in series with RG2 (RSERIES- ) to cancel the input offset voltage.

Summary and Proof
To summarize, ideal performance can be obtained from input filters with series resistors between the sense resistor and RS+ and RS- pins if:
1. The series resistor between R SENSE+ and RS+ is kept small with respect to RG1
2. The series resistor between R SENSE+ and RS+ is twice as big as that between RSENSE- and RS-

Note, finally, that since RSERIES+ is twice RSERIES- , the common-mode filter capacitors will also need to be suitably scaled to meet desired AC and transient performance objectives.

The bench-test results in Table 1 were obtained with MAX4173T and support the discussion above. The min and max values of VOS were calculated using min and max bias currents from the data sheet; the min and max gain errors were calculated using ±30% of RG1 = 6 kO.


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Table 1: Series resistor test results for the MAX4173.

Similarly, bench results obtained with MAX4372F are shown below in Table 2 (RG1 = 100 kO).


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Table 2: Series resistor test results for the MAX4372.

The derivation of calculated minimum and maximum gain errors and min-max VOS is:

Old Gain
= Constant × RGD / RG1
= 20 (for T-version of MAX4173)

New Gain
= Constant × RGD / RG1 new; RG1 new = RG1 + (RSERIES+ )
= Old Gain × RGD1 /RG1 new
= 20 × RGD1 /(RG1 + RSERIES+ )

Gain Error
=(20-NewGain)/20%
= RSERIES+ /(R G1 + RSERIES+ )

Minimum Gain Error
= RSERIES+ /(1.3 × R G1 + RSERIES+ )

Maximum Gain Error
= RSERIES+ / (0.7 × R G1 + RSERIES+ )

RG1 = 6 kO for MAX4173

VOS
= IBIAS1 × R G1 ” I BIAS2 × RG2
= I BIAS1 × (R G1 ” 2 × RG2 ); where I BIAS2 = 2 x I BIAS1

Ibias1 (min) = 0
Ibias1 (max) = 50 µA for MAX4173

About the authors

Prashanth Holenarsipur is a corporate applications engineer and Houston Hoffman was an intern, both at Maxim Integrated Products, Inc., Sunnyvale, CA.

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